Basic Type

The receivers used in sonar are generally of the conventional superheterodyne type with a few added circuits that are peculiar to sonar.

The receiving systems used with the scanning type of echo-ranging equipments are of the dual-channel type, which is required in present methods of video portrayal. Deviation indicators, which are cathode-ray tubes or galvanometers, use either a sum-and-difference or a comparison receiving system. These systems are used to measure the phase angle of the echo signal between the two halves of a transducer that has been electrically split so that, on reception, it acts as two independent hydrophones. When the proper circuits are used with these systems the phase angle can be translated into voltage differences, and the video portrayal is indicative of the deviation from the correct target bearing, that isodepression deviation indication (DDI) or bearing deviation indication (BDI). Scanning systems use conventional superheterodyne receivers-one for the video channel and one for the audio channel. The only function of the receiver in the video portrayal is to furnish brightening voltage to the grid of the cathode-ray tube because the scanning and deflection voltages are developed outside the receiver circuits. The audio channels are used to supply the returning echoes to the operator as an aid in identifying targets.

  The transducer used with scanning systems is keg-shaped and is mechanically and electrically divided into an even number of independent elements. In the model QHB-1 there are 48 such elements located so that each element covers an arc of 7 ½° of the transducer's periphery. On transmission, the keying relay connects all the elements in parallel so that sound power is radiated in all directions simultaneously, whereas on reception the elements are connected so as to form a sharp beam in the horizontal plane. The output of each element is connected through its individual preamplifier to its corresponding element on each of the two scanning switches.

The video scanning switch is driven at a constant rate and has a control transformer geared at a 1-to-1 ratio with it. The output of this transformer controls the positioning of the electron beam of the cathode-ray indicator so that the electron beam synchronizes in azimuth bearing with the scanning switch.

The audio scanning switch and the video switch are mechanically and electrically identical, but they differ in application. The audio switch is positioned by a servo system and must be manually trained to the desired bearing.

These scanning circuits are called the directional sensitivity circuits. They were discussed in chapter 6.

Conventional Superheterodyne Sonar Receiving System
The receiving system described in the following paragraphs is that of the QHB-1 scanning sonar equipment. The QHB-1 system was chosen for discussion because its method of video portrayal follows the conventional design and it can be considered a typical system. The block diagram is shown in figure 7-1.   The receiver-converter includes separate video-channel and audio-channel receivers for the signals from the corresponding scanning switches. In this system a tunable oscillator supplies a frequency to the first mixers in both channels and also to the converter, which produces the transmitted frequency. This oscillator identifies the

Block diagram of QHB-1 dual-channel receiver unit.
Figure 7-1. -Block diagram of QHB-1 dual-channel receiver unit.
circuit as the unicontrol system because it enables a single control to tune the receivers and the transmitter at the same frequency. In the receiver-converter a master fixed-frequency 65 kc oscillator (not shown in figure 7-1) modulates a 90.5 kc   signal (arbitrarily selected) from the unicontrol oscillator, and the 25.5 kc frequency difference is amplified in the transmitter power amplifier. The i-f stages of both receiver channels are tuned to 65 kc. The first mixer in the receiver channels

automatically produces an intermediate frequency of 65 kc because the 25.5 kc received signals are mixed with the 90.5 kc output of the unicontrol oscillator.

The frequency of the unicontrol oscillator may be varied from 87 to 94 kc to produce a variation in transmitted frequency of from 22 to 29 kc. The transducer is operable between 24 and 27 kc, Although its most efficient operating point is at its own resonant frequency-in this case, 25.5 kc. The operation of the unicontrol oscillator will be studied with the receiver circuits; the details of the master oscillator will be studied in the chapter on transmitter circuits.

A bias for the reverberation control of gain (RCG) derived from the audio output is supplied to the two r-f stages of the audio channel and to the two r-f stages of the video channel. In addition, the master gain control on the front panel of the sonar indicator operates upon the two r-f stages of each channel.

In the audio-channel receiver a beat-frequency oscillator is controlled, except in the listen condition, by two types of doppler-nullifier circuits-own-doppler nullifier (ODN) and target-doppler nullifier (TDN). The ODN circuit is operated by ship's own doppler and restores the reverberations to 800 cycles per second; the TDN is operated by target echo signals and has such short response time that it restores the audio note on target echoes to 800 cycles per second. The doppler-nullifier circuits are necessitated by the sharply tuned audio circuit in this receiver. The TDN circuit can be disabled and is used only when the audio system is operated in the peak filter position for improvement of the signal-to-noise ratio.


The video channel of the receiving system (figure 7-2) consists of a tuned r-f stage, V701; and untuned r-f stage, V702; a mixer, V703; an i-f amplifier, V704; and an output rectifier and cathode follower, V705.

The balanced input signal is supplied from the video channel of the directional-sensitivity circuits to the input transformer of the tuned r-f stage. From this transformer it is coupled through a suppressor resistor to the t-r-f amplifier tube.

The RCG bias for the grid circuit, derived as explained later in this chapter, is supplied to the

  control grid of this stage. The cathode bias is controlled by the master gain control on the front panel of the sonar indicator control. The plate load of the stage consists of an inductor and three capacitors in parallel. One of these capacitors is a section of the master tuning capacitor. The other two ensure correct tracking of the circuit with the audio t-r-f stage and unicontrol oscillator.

The voltage output across this tuned impedance is capacitively coupled to the grid of the untuned r-f stage. RCG grid bias is applied also to this stage, as is the cathode bias from the master gain control. This r-f stage uses degenerative feedback provided by an unbypassed cathode resistor. The output of this stage is capacitively coupled to the control grid of the mixer. This stage operates at almost zero d-c grid bias, and uses a conventional cathode resistor for inverse feedback. The screen voltage is produced by the cathode-follower section of the unicontrol oscillator, and consists of a d-c component and an a-c component at the unicontrol oscillator frequency.

The i-f stage is conventional except that L-C coupling is used rather than the standard i-f transformer coupling. The plate and grid inductors are tuned to parallel resonance at the 65-kc intermediate frequency. R714 serves as a parasitic suppressor.

Inverse feedback is obtained in the i-f stage across the cathode resistor. This stage has well-decoupled plate voltage and is followed by a band pass i-f coupling, similar to the preceding one. The coupling capacitor, in conjunction with the tuned i-f transformers, serves to form a band-pass filter.

The output of the i-f stage is rectified by one-half of the twin triode, V705, connected as a diode, and the output of this rectifier appears across a resistive load with the i-f frequency component filtered out. This output is produced in series with a fixed negative bias of 10 volts which is also impressed upon the grid of the cathode-follower section of V705. The rectified signal is coupled to this grid through an RC circuit with a time constant of 22 milliseconds, which is adequate for pulse reception yet sufficiently short for removing any steady-signal (low-frequency) components. The capacitor of this RC circuit must, of course, be shorted whenever test measurements of steady-signal levels are to be made in the


video-channel output. The cathode-follower section of V705 produces its output across three parallel resistors for transmission to the sonar indicator control and for utilization in the video display. Feedback caused by the impedance of the positive power supply which is common to the entire video channel is prevented by a decoupling filter consisting of two parallel resistors in series with the supply and a capacitor to ground.


The audio channel of the receiving system consists of a tuned r-f stage, V706; an untuned r-f stage, V707; a first mixer, V708; a second mixer, V709; an audio amplifier, V710; and an audio power output stage, V711.

The stages ahead of the second mixer are identical with those of the video channel, and the discussion of the video stages is applicable also to the audio circuits. The only differences are an additional adjustable gain control in the audio circuit, and, naturally, the circuit symbol numbers.

The i-f circuit coupling between V708 and V709 is the same as in the video channel except for the value of the coupling capacitor and suppressor resistor. The output of the i-f stage is supplied to the second mixer, V709, through a current-limiting resistor. Both V708 and V709 use inverse feedback across their cathode bias resistors. The d-c component of the screen voltage is supplied by a divider from B+ and the a-c component is supplied from a tap in the beat-frequency-oscillator (BFO) circuit.

The a-f component in the anode output of the second mixer, V709, is capacitively coupled to the grids of audio amplifier V710. The plate circuit of the second mixer, V709, contains the proper R and C components to ensure a broad audio response in this channel. The two sections of the twin triode, V710, are used as separate audio amplifiers, with common cathode bias. The grid signals are identical for both sections, and the output across the plate resistor of section 1 is used for the doppler-nullifier, whereas the output from section 2 of the tube is used for the audio signal channel. At this point, the audio channel is provided with a peak filter, which consists of an inductor and a capacitor tuned to 800 cycles per second. This filter serves as a load on

  the triode section of V710, and produces a voltage 6 db below the peak value for frequencies 50 cycles per second above or below the center frequency. The filter operates only when the doppler-nullifier-transfer relay is energized by the audio peak-band switch on the control console.

The audio voltage at this point is supplied to the grid of the output amplifier, V711, by a divider and filter network, which further discriminates against high frequencies.

The audio power output amplifier, V711, is a conventional beam-power tube. This stage is transformer-coupled to a 250-ohm line for operation of speakers and chemical recorders. The two capacitors across the primary of this transformer provide additional attenuation of the high frequencies.


The RCG and TVG (time varied gain) circuits are arranged to control the grid bias of the r-f stages in the receiver channels and thus to reduce the receiver sensitivity during transmission and heavy-reverberation periods. A reverberation-controlled gain in the audio channel restores the output as the reverberation level decreases along the particular bearing to which the audio system is trained. The gain recovery of the video channel is identical with the audio channel and employs the RCG voltage. The circuits producing this voltage use V712 and one section of twin triode, V713.

During the interval preceding a transmission pulse, the control grid of V713 is maintained at approximately -80 volts, with respect to ground, by a divider connected between the keying pulse line and the o105 volt bias. The cathode is connected to approximately a -60 volt point on another dividing network connected to the same -105 volt source. In this condition the grid bias is maintained at cut-off. The keying pulse changes the voltage between grid and ground raising the grid voltage above cut-off. This rise in grid bias in a positive direction allows the tube to conduct, thus charging the RCG bias capacitor, C715, negatively to ground. At the end of the keying pulse, the grid of the TVG control tube, V713, is restored to approximately -80 volts with respect to ground and the tube no longer conducts.


FOLDOUT - Figure 7-2. -Dual-channel receiver circuits.

The RCG bias capacitor can then discharge through the resistors shunted across it and through half of diode, V712. The time constant is such that the voltage of the capacitor, if used for grid bias in the gain-control tubes of the receiver, restores the gain to within 5 db of its original value at a time corresponding to a range of 80 yards, provided there is no d-c voltage across the other half of the diode, as would normally be produced by an audio output signal. This voltage and gain recovery is defined as TVG and is illustrated in figure 7-3.

TVG and RCG bias.
Figure 7-3 -TVG and RCG bias.

Reverberation control of gain is affected by retarding this recovery as a function of audio output level. The primary voltage of the audio output transformer is coupled to one cathode of the duodiode, V712, through a capacitor and current-limiting resistor. Rectification by this diode produces a d-c voltage across the resistor R755 of the diode proportional to the audio level. This voltage is applied to the second half of V712 through R756 and causes the discharge of C715 to be retarded. The discharge of the RCG bias capacitor, C715, through the second diode section of the RCG control tube, V712, can proceed only when the potential on C714 is less than that on C715. As the reverberation and thus the d-c voltage across the capacitor C714 decrease, the RCG bias capacitor can discharge further, allowing the gain to increase. An increase in reverberation level cannot, however, decrease the gain. When the reverberation level is high and persistent, an overriding control is provided by the shunting action of the 2.2-meg resistor across the RCG bias capacitor. This shunt resistor allows the capacitor to discharge independently at a slow rate, whether or not the audio level allows it to

  discharge through the diode. This RCG voltage (figure 7-3) is supplied to the grid circuits of the r-f stages of the audio and video channels.

The same RCG bias that is supplied to the audio channel is used in the video channel. Because it is not desirable for the gain of the video and the gain of the audio channel to depart from some desired ratio one to the other. Otherwise, it might cause an absence of audio or video signals, depending on which gain is lower. Under normal conditions the reverberation is sufficiently omnidirectional to produce satisfactory operation.


The unicontrol oscillator (figure 7-2), tunable from 87 kc to 94 kc supplies the screens of the first mixer tubes in the video and audio channels of the receiver. It also supplies the converter circuit for producing the frequency to be transmitted. The unicontrol oscillator, half of V714, is a Hartley oscillator. The oscillator employs a center-tapped inductor, tuned by four capacitors and a section of the main tuning capacitor. A value of inductance was chosen to provide good tracking with the t-r-f stages of the receiver. The padding capacitance, C770 and C795, helps to provide tracking with the t-r-f circuits. The remaining two capacitors C769 and C768 are a trimmer and fixed capacitor.

The voltage from the center tap of the inductor of the Hartley oscillator is supplied to the high-impedance grid circuit in the converter by means of a series resistor and capacitor. This same voltage is supplied to the control grid of the second triode section of V714. This section (a cathode-follower) provides screen grid modulation to the first mixer tubes, V703 and V708, in the video and audio channels, respectively. Since the screen grid voltage is supplied from the large unbypassed cathode resistor of the cathode-follower the screen voltage will be very low during the negative half of the input signal to the cathode-follower. The result is a screen voltage that is low for an appreciable part of each cycle. This mode of operation results in improved mixer action and provides better discrimination against undesired frequencies.


The beat-frequency-oscillator (BFO) circuit, figure 7-2, consisting of one section of twin triode V715 and reactance tubes V716 and V717,


modulates the screen voltage of the second mixer, V709, in the audio channel of the receiver in order to produce an audio note from the received signals. The oscillator frequency is controlled by the target-doppler-nullifier and own-doppler-nullifier control tubes, in response to the functioning of the doppler-nullifier circuits. The oscillator consists of a Hartley circuit with grid leak bias and cathode degeneration for stability. The oscillator coil assembly is an arrangement of three inductors and two blocking capacitors, which provide (1) a center tap for the oscillator cathode and (2) d-c isolation for the anodes of the reactance tubes. The center tapped inductor functions in parallel with tuning capacitor C775 and is effectively in parallel with the plate circuits of the reactance tubes.

A resistance of 50 ohms, composed of two resistors, R778 and R781 in parallel, in series with capacitor C774 is used as a phase-shifting circuit. The voltage across R778 and R781 leads the output voltage of the oscillator circuit by nearly 90° and is the a-c grid signal for the reactance tubes. Each of these tubes has suppressor resistors in series with its grid to prevent spurious oscillations. Both tubes are cathode-biased. The screen voltage comes from the regulated 150-volt supply. The reactance tube, V716, controls the BFO in response to the ODN circuits, whereas tube V717 exerts control from the TDN circuits.

The functioning of the circuit in response to the d-c controls can be analyzed in the following manner. Let e represent the a-c voltage developed across the phase-shifting resistors R778 and R781, and supplied to the reactance-tube grids. As ip is in phase with e, the anode current that these tubes draw from the oscillator circuit leads the output voltage of the oscillator circuit by 90° and is thus the equivalent of the current in an adjustable capacitor. The adjustment results from changes in ip, caused by changes of the d-c grid voltage. A change of this voltage in the positive direction increases ip, that is, effectively the capacitance has increased. This lowers the frequency of the oscillator. Similarly a change of d-c grid voltage in the negative direction increases the frequency.

A voltage, taken from the center tap 3, of the lower inductor supplies the a-c component of screen voltage through R744 to the second mixer, V709, in the audio channel.


The doppler-nullifier circuits, which provide the d-c control voltages for the BFO, consist of an audio amplifier and limiter, a power amplifier stage, V719, a discriminator, and relay control circuits.

Audio Amplifier and Limiter

The audio frequency voltage, developed at the anode of half of the first audio amplifier, V710, is coupled to potentiometer R810 (labeled "D. N. GAIN") for the purpose of adjusting the amplification of the audio circuit to a value adequate for doppler nullification. The signal from the potentiometer is coupled to the grid of V718 through a. low-pass RC filter to reduce any 65-kc signal component present in the audio channel. The series resistor R811 limits the input grid signal on the positive half cycle. The increase in plate current at this time is further limited by the degenerative action of the unbypassed screen voltage supply. The cathode bias is such as to limit the plate current on the negative portion of the input signal. Thus essentially a square wave output of approximately constant amplitude is produced from V718. This is desirable because the input to a frequency discriminator should vary only in frequency.

The square-wave output is coupled to the control grid of beam-power tube V719 capacitively and through a current-limiting resistor. Because of the necessity for preserving the square-wave character in the incoming signal, the power stage employs negative feedback provided by the unbypassed screen supply and cathode resistor. Capacitor C776 and five series resistors are connected across the primary of the output transformer to reduce the shunting effect of the primary inductance by making it part of a low-Q parallel-resonant circuit in the vicinity of the 800-cps frequency. The result is a voltage from the secondary of T705, which approximates a constant amplitude square wave which is coupled to the discriminator.

To make the doppler-nullifier circuits inoperative when the equipment is in the listen condition, one-half of twin triode V715 is used to cut off the beam-power tube, V719. In the listen condition, the keying selector on the front panel of the


sonar-indicator-control unit, connects the keying pulse line to a circuit of -105 volts with respect to ground. Conduction occurs in V715 which effectively connects a voltage divider circuit comprising V715 and resistors R816 and R817 from the keying pulse line to ground. This action biases the grid of the beam-power tube to cut-off. The need for a zero output is explained in connection with the ODN operation in the following paragraphs. When the equipment is echo ranging, this keying pulse line is normally at +45 volts above ground, and it pulses to +220 volts during transmission. Under these conditions, the diode-connected half of V715 cannot conduct and it has no effect on the operation of power amplifier V719.   Discriminator Circuit

The function of the discriminator circuit is to produce a d-c voltage that is proportional to the deviation of the audio frequency from the reference of 800 cycles per second. This function is accomplished by a circuit that produces voltages whose amplitude is proportional to the variation in frequency and a comparison rectifier with a filtered output.

The discriminator circuit (figure 7-4) functions in a manner somewhat similar to the ratio detector used with f-m circuits at higher radio frequencies. For the discriminator circuit of figure 7-4, the d-c voltage produced by rectifier V720 must be

Figure 7-4 -Discriminator. A, Equivalent circuit; B, voltage curve; C, output curve.

positive for audio frequencies above 800 cycles per second and negative for frequencies below this value; and there must be no secondary crossover points due to harmonics of frequencies which are below 800 cycles per second. The input impedance of the circuit shown in figure 7-4, A, is constant and equal to R/2 over the range of frequencies involved. It is therefore a desirable load.

The voltage across the parallel combination of resistance and capacitance is shown as curve eAO in figure 7-4, B, whereas that across the resistance-inductance combination is shown as curve eBO in the same figure. These two voltages are equal to each other at a frequency such that the inductive and capacitive reactances are equal. The ratio of these two voltages is approximately 4-to-1 at the frequencies which are either one-half or twice the center frequency. One of the two desired output voltages is produced to ground across capacitor C717 and the other across inductor L713.

These two voltages, eAO and eBO are rectified by the two sections of twin diode V720. Resistors are placed in series with the rectifier loads to produce d-c voltages that are proportional to the average value of the a-c voltages, rather than to the peak values, to minimize the effects of waveform distortion. The a-c voltage across the capacitor varies inversely with frequency while that across the inductor varies directly with the frequency. Thus, the a-c voltage that varies directly with frequency produces a positive d-c voltage output from V720, while the a-c voltage that varies inversely with frequency produces a negative d-c voltage output. These voltages are developed across separate RC filter combinations. The difference between these two d-c voltages is developed across three resistors in series connected across the two filters mentioned. A close adjustment of the exact frequency that produces zero d-c voltage is made possible by the center resistor of the three, which is a potentiometer, so that this zero-voltage point can be made to coincide exactly with the center frequency of the audio peak filter in the audio channel.

A typical d-c voltage curve from the discriminator circuit is shown in figure 7-4, C. For frequencies above 800 cycles per second this voltage is positive, and for frequencies below 800

  cycles per second it is negative. The value of this voltage decreases as zero frequency is approached because the audio response falls off at very low frequencies. At zero frequency, there is a small residual voltage that is due to power-supply ripple. The discriminator output voltage is supplied to relays K702 and K703 for use in the ODN and TDN operations.

Own-Doppler-Nullifier Circuit

The combination of the BFO with reactance tubes and of the audio amplifier with discriminator circuits provides a means of making the audio frequency correct itself to the reference frequency of 800 cycles per second. It is similar to a servo system in which the audio-frequency input deviation produces a d-c voltage output that helps to restore the frequency of the input signal to normal. The total feedback is about 30, which means that the audio frequency deviation is reduced to one-thirtieth of that which would be obtained without doppler nullification.

For ship's own doppler nullification (ODN) the circuits function in the following manner. During the interval between transmission pulses, the keying pulse line remains at +45 volts to ground. The grid, terminal 4, of V713 is held at its cathode potential (approximately +5 volts to ground) by grid current and C723 charges to about 40 volts. During transmission when the keying pulse line rises to +240 volts for a period of 35 milliseconds, capacitor C723 in the grid circuit of the second section of V713 is charged by grid current to approximately +235 volts. During the period of this pulse, the ODN sampling relay is not disturbed, but remains closed-its normal position-for its current change is very slight. At the end of the transmission, when the keying pulse line is restored to +45 volts, the grid of the tube is carried to approximately -190 volts with respect to ground by the charge retained on the grid capacitor. This action cuts the tube off, and the sampling relay, K702, drops out. (Capacitor C726 is shunted across second section of V713 to reduce the otherwise severe voltage transients.) The grid capacitor C723 discharges towards +40 volts, and as the grid potential approaches +5 volts with respect to ground a point is reached at which the tube current is again sufficient to operate the sampling relay, which remains energized until


the end of the next keying pulse. During the time this relay is de-energized, it applies the output of the discriminator to the ODN reactance tube, V716, as a d-c grid bias.

During the sampling period, which is equivalent to approximately 250 yards of range, capacitor C710 in the grid circuit of reactance tube V716 acquires a charge from the frequency-discriminator circuit. Resistor R858 placed in series with the charge path of the capacitor sufficiently lengthens the time constant to enable the circuit to average the frequencies present in the reverberation and to approach a true solution for own ship's doppler. When the sampling relay closes at the end of its period, capacitor C710 retains a charge which is applied to the grid of the reactance tube thus maintaining the audio note at 800 cycles per second. This frequency can shift only as a result of changes in the charge on the grid capacitor caused by leakage.

During the sampling period, contacts 4 and 5 on the ODN sampling relay ground capacitor C709 in the grid of TDN reactance tube V717 through R867 provided that the doppler-nullifier relay is energized. This relay will be energized when the equipment is operating in the audio peak condition and the TDN switch is on. The purpose of this operation is to remove any residual charge on capacitor C709 that results from the TDN operation and thus to provide the correct initial reference for establishing the ODN circuit control. After the sampling period, the doppler-nullifier relay reconnects the TDN circuit to the reactance tube so that it may function for the rest of the time, if so desired.

When the equipment is operated in listen condition, the keying pulse line is connected to -105 volts, so that the second section of V713 remains cut off continuously, and the ODN sampling relay remains open. The -105 volts on the keying pulse line cuts off the input to the discriminator

  as mentioned previously through the action of the diode-connected half of V715 with the result that the BFO remains stable during the listening operation.

Target-Doppler-Nullifier Circuit

As a means of improving the signal-to-noise ratio, this receiver can be operated with an audio peak filter. The audio peak filter comprises a parallel resonant circuit shunted across the output of the second triode section of V710. Because this filter removes off-frequency signals resulting from target doppler, it is necessary to accompany its use with a TDN circuit. This circuit functions in the same manner as the ODN circuit and uses the same audio amplifier and discriminator to restore the audio frequency to 800 cycles per second for all signals.

When the switch on the sonar indicator-control is thrown to audio peak, it energizes the doppler-nullifier relay, if the equipment is echo ranging, but not if it is listening. This relay has two functions-(1) it grounds one terminal of the audio peak filter, making it operative, and (2) it connects the discriminator output to the grid reference capacitor C709 of the TDN reactance tube through the sampling relay when the latter is energized. During the reverberation sampling immediately after transmission, before the sampling relay closes, the discriminator is connected to the ODN reactance tube in order to set the BFO for an audio output of 800 cycles per second from the reverberation. At the end of this period, when the sampling relay recloses, the discriminator output is applied to the grid circuit of the TDN reactance tube, and all signals thereafter are restored to the same reference. The TDN circuit can be disabled by closing switch 5701, which allows (1) echo ranging on fixed targets with the audio peak filter and (2) the use of the ODN circuit alone.

Sum-and-Difference Receiving System
To convert the echo-signal phase-angle difference between the two halves of a split transducer into a polarized magnitude difference, either a sum-and-difference or a comparison system must be used. In operation, there is very little difference between the results of the two systems, and their complexity is about the same. The receiver


  described here uses a sum-and-difference system and is part of the Model QDA depth-determining equipment.


In the discussion of this receiver, when the echo is returned from a target below the axis of the


Block diagram of the QDA receiver.
Figure 7-5. -Block diagram of the QDA receiver.
transducer, a down voltage is developed. This down voltage indicates that the phase of the signal from the lower half of the transducer is leading that of the upper half, and it is necessary for the operator to train downward to obtain the correct depression angle.

Similarly, when the echo is returned from a target above the axis of the transducer, an up voltage is developed, indicating that the phase of

  the signal from the upper half of the transducer is leading that from the lower half. In this case it would be necessary for the operator to train upward for the correct depression angle.

The circuits that produce the up and down voltages, together with the indicating unit, are called the depth deviation indicator (DDI).

In azimuth equipments, operation is very similar, except that the plane of operation is rotated


Hybrid-coil input circuit.
Figure 7-6 -Hybrid-coil input circuit.

90°, and down and up become right and left. The unit is then called the bearing deviation indicator (BDI).

Figure 7-5 is a block diagram of the QDA receiver. The voltages from the two halves of the transducer are combined in a hybrid coil. This coil has two output voltages, one of them being the sum of the up and down voltages, the other being the difference or diff, as it will be referred to, of the up and down voltages. These two output voltages are then amplified in separate receiver channels. The two channels are very similar, each consisting of two stages of r-f amplification, a modulator, and a 2-stage i-f amplifier. The two channels are controlled independently by means of two RCG circuits.

Because of the use of RCG circuits, the reverberation outputs of the sum and diff channels are substantially constant. Thus a d-c bias may be introduced in series with the conjugate detector of the proper magnitude to prevent the reverberation noises from appearing at the output. However, any noise or signal of greater magnitude than the average reverberation level appears on the deviation indicator scope as a distinct pulse. This bias is known as threshold control and is manually controlled by a potentiometer located on the receiver chassis. With this control, the threshold voltage may be set as close to the average reverberation level as desired, or it may be removed from the circuit entirely.

  The output of the conjugate detector is supplied to the vertical deflection coils of the DDI through the vertical deflection amplifier, which converts the positive and negative d-c signals of the detector into the proper voltage for the operation of the cathode-ray tube.

The horizontal-sweep generator in the block diagram of figure 7-5, controls the left-to-right traverse of the electron beam of the DDI cathode-ray tube. The sweep progresses from left to right at a nonlinear rate. The nonlinearity of the sweep is evidenced by the fact that during the first half of the sweep the beam travels a time equivalent to 500 yards of sonar range, whereas during the second half of the sweep the sweep requires twice that time, or the equivalent of 1,000 yards of range, to travel the same distance.

The hybrid coil used in this receiver is shown in figure 7-6. The signals from the separate halves of the transducer are connected at the inputs marked "E." The signal current in the primary windings 4-5 and 5-6 of the sum transformer is proportional to the vector sum of the up transducer voltage, and the down transducer voltage. The sum may be checked by observing the instantaneous polarity markings at the two sources and the direction of the arrows in the figure. Thus, the voltages induced in the two sections of the primary winding of the sum transformer combine to produce an output which is proportional to the vector sum of the voltages from the two halves of the transducer.

If the transducer voltages are equal and in phase-a condition which would exist when the

Sum and diff voltages related to up and down
Figure 7-7 -Sum and diff voltages related to up and down phases.


target is exactly on the axis of the transducer-the up and down voltages are in series and in phase so that the voltage on the low side of the sum transformer is represented by 2E. If the ratio of the transformer is N for the secondary winding (S1 and S2 in series) to the four primary windings (P1-P4 in series), the secondary voltage for the on-target case will be 2NE.

For the on-target case the potential difference between points 2 and 5 is zero, because the two transducer voltages, being of equal magnitude are opposite in phase and subtractive in their effects across these points. Under these conditions no current flows through the diff half of the transformer because this half is connected across points 2 and 5 and the output of the diff channel is zero.

If the voltages from the two halves of the transducer are equal and 180° out of phase, the output of the sum channel will be zero, and the output of the diff channel will be 2NE, which is the same as the output voltage of T701 for the on-target case.

If the target is above the axis of the transducer the signal coming to the upper half of the transducer arrives before that coming to the lower half, resulting in a phase differential between the two signals. The relative magnitudes of the two signals is approximately the same, because the difference in range between the upper and lower half of the projector is a very small percentage of the total range to the target. If the target is above the axis of the transducer, the signal supplied to the up input of the hybrid transformer leads that supplied to the down input, and results in a potential difference between points 2 and 5 of figure 7-6. Thus, at that instant, 5 is negative and 2 is positive, the direction of instantaneous current flow is indicated by arrow A, for the polarities shown, and results in an output from the diff channel, which is proportional to the phase difference between the up and down signals.

If the target is below the axis of the sound beam, the signal supplied to the down input is leading the signal supplied to the up input. This results in a 180° phase shift from the preceding case. Thus, at the instant that point 2 is negative, point 5 is positive and the current flows in the direction indicated by arrow B. Note the direction is opposite to that existing when the up input was leading. Thus the difference signal when the up

  Vector diagrams of input voltages.
Figure 7-8 -Vector diagrams of input voltages.

signal is leading, is displaced 180° with respect to its phase when the down signal is leading. Figure 7-7 shows a plot of the amplitude of the sum and diff voltages as a function of the phase angle between the up and down voltages.

Figure 7-8 shows that when the target is above the transducer axis the sum voltage lags the diff voltage by 90° and when the target is below the axis the sum voltage leads the diff voltage by 90°. It is not possible algebraically to add two voltages in quadrature. In order to overcome this difficulty the diff voltage is shifted 90°. The solid curves of figure 7-7 show the relative magnitudes of the sum and diff voltages. These curves are not coincident in time. The dotted curve with the solid curve labeled "diff" in the figure shows the diff voltage after the phase has been shifted 90°. This shifted voltage can now be added directly to the sum voltage in the conjugate detector.

Consider the instantaneous signal at the up and down hydrophones of some particular amplitude, E. For an on-target signal, the two voltages are in phase. The diff voltage is zero and the sum voltage is 2E. If the echo originates above the axis of the transducer, up leads down, but if the echo originates below the axis of the transducer, down will lead up.

Figure 7-8, A, shows a vector diagram for an on-target signal. Suppose the up voltage leads

  the down voltage by an angle θ. Figure 7-8, B, illustrates this case for θ equal to 20 electrical degrees, or about 0.4 mechanical degree. Because the vectors are equal in magnitude, it follows from the geometrical construction of sum and diff, that for any up signal the diff leads the sum by 90°. Similarly, figure 7-8, C, illustrates that if the up voltage lags the down voltage by an angle θ, the diff voltage lags the sum voltage by an angle of 90°. In figure 7-8, B and C, it is shown that the phase of the diff voltage with respect to the sum voltage changes by. 180° as the returned echo changes from up to down. The conjugate detector uses this fact to determine whether the trace of the DDI scope is deflected up or down.

Figure 7-8, D, illustrates the difference of 180 electrical degrees between the up and down voltages produced by a target that is approximately 4 mechanical degrees above the axis of the transducer. This case is similar to the on-target case except that now the sum is zero and the diff is equal to twice either the up or the down voltage.

From the foregoing discussions and the geometry of figure 7-8 it can then be concluded that:

1. The amplitude of the sum voltage is equal to 2E cos θ/2, where θ is the electrical phase difference between up and down.

R-f amplifier.
Figure 7-9 -R-f amplifier.

Typical frequency characteristics, input circuit,
and r-f amplifier.
Figure 7-10 -Typical frequency characteristics, input circuit, and r-f amplifier.

2. The amplitude of the diff voltage is equal to 2E sin θ/2.

3. The sum and diff voltages are always 90° apart.

4. As the diff voltage goes through zero amplitude, at zero phase difference between up and down, the phase of the diff voltage changes by 180°.

5. If the up voltage leads the down voltage, the diff voltage leads the sum voltage by 90°.

6. If the up voltage lags the down voltage, the diff voltage lags the sum voltage by 90°.


The r-f amplifier is shown in figure 7-9. It is a typical 2-stage transformer-coupled amplifier. The input transformer has sufficient voltage gain to reduce the effects of inherent tube noise. The secondaries of the coupling transformers form band-pass circuits resonated at approximately 55 kc. The capacitance includes that which is inherent in the secondaries, plus 10 μμf of shunt capacitance in parallel with the input capacitance of the tube and circuit wiring. Resistors are shunted across the primaries of T702 and T703 to prevent excessive gain and broaden the resonance curves of the amplifiers.

Grid bias is obtained from the RCG circuits, which will be discussed later in the text.

A typical gain-versus-frequency characteristic curve of the input circuit and r-f amplifier is shown in figure 7-10.


It was previously explained that a phase difference of 90° exists between the sum and diff

  voltages. For proper operation of the conjugate detector, which will be explained later, a differential of zero or 180° must exist between these voltages for the algebraic combination of them. It follows, then, that the phase of either the sum or diff channel must be advanced or retarded by 90° to fulfill the zero or 180° relationship.

A phase-shifting network could be placed directly in the sum or diff channels and effect the necessary change in phase. However, unless they are very complex, most phase-shifting networks are sensitive to changes in frequency. The incoming signal is composed of the transmitter frequency, the necessary sidebands to produce the pulse modulation of the carrier, and the 800-cps modulation frequency, which is used to produce an audible signal in the audio circuits after detection. A phase shifter which would encompass this wide range of frequencies would be impractical because of its complexity. If the network could be allowed to shift the phase of a single frequency a much simpler circuit could be devised.

Like all superheterodyne receivers, this receiver uses an r-f signal from a local oscillator for frequency conversion. In this case the signal is from the unicontrol oscillator, and the frequency is from 240-250 kc. This signal, of course, consists of a single frequency.

A review of the theory of modulation will show that if the frequency of one of the modulation

Modulator schematic.
Figure 7-11 -Modulator.


I-F amplifier schematic.
Figure 7-12. -I-F amplifier.
signals is changed in phase, the resultant frequencies will be changed in phase by that same amount. The derivation of this principle is complex and is not discussed here.

From the block diagram (figure 7-5) it can be seen that there is a phase shifter in the circuit supplying the unicontrol oscillator signal to the diff channel. This network, in effect, retards the diff signal by 90°, without affecting the sum signal, which will provide the necessary 180° between the sum and diff channels.

Then, the sum and diff are in phase for any up signal, and the amplitude of sum+diff is simply the sum of the amplitudes. For any down signal the two voltages are 180° out of phase, and sum+diff is the difference between the amplitudes of the voltages. Figure 7-7 shows the relation of the sum and diff voltages to the phase difference of the up and down signals, after the 180° phase shift. Curve D, plus the right half of the solid diff curve, represents the diff voltage after the phase shifter effect has been introduced.

The modulator circuit is shown in figure 7-11. The function of the modulator is to convert the input frequency-which is between 50 and 60 kc-to a fixed frequency of 190 kc by modulating the input with a variable frequency of from 240 to

  250 kc. The 240-to-250 kc oscillator is always related to the 50-to-60 kc input frequency by a fixed difference of about 190 kc. This difference is maintained in the oscillator unit by the unicontrol system.

The modulator is a push-pull circuit for the input frequency of from 50 to 60 kc-that is, the signals on the grids are 180° out of phase. The 240-250 kc signal introduced between the center tap of the input transformer and ground makes the 240-250 kc signals on the two grids in phase and this frequency is suppressed in the output. The sum and diff frequencies are obtained most effectively by biasing the two tubes nearly to cut-off on no signal.

The principal output frequencies are the input frequency (50 to 60 kc), the sum frequency (290 to 310 kc), and the difference frequency (190 kc). The input and sum frequencies, as well as products of higher order, are suppressed in the output by the plate load impedance, which acts as a band-pass filter that passes frequencies in the 188.4-to-191.6 kc band.


A schematic of the i-f amplifiers is shown in figure 7-12. There are two of these amplifiers and


they are identical-one is used in the sum channel and the other in the diff channel. The coupling transformers are of conventional type, having both the primary and secondary tuned.

The first stage is supplied with RCG bias, but the second stage is not. The second stage derives its bias from two sources-(1) from a voltage divider between -105 volts and ground supplying -18.5 volts to the grid of the stage and (2) from a variable resistor inserted in the cathode return. This resistor is marked "sum gain" in the sum channel and "diff gain" in the diff channel. The purpose is to provide a means of compensation for differences in gain between the sum and diff channels. The range of control of each channel is about 12 db.

Frequency characteristics of the i-f amplifier.
Figure 7-13 -Frequency characteristics of the i-f amplifier.

I-F Bandwidth-Doppler

The bandwidth should be just wide enough to pass the frequencies necessary for proper performance of the system. If the band were wider the signal-to-noise ratio would be lowered, with a consequent loss of weak signals.

The transmitter frequency is 55 kc and is amplitude modulated with an 800-cps signal. The three principal frequencies transmitted into the water are 54.2, 55.0, and 55.8 kc. Therefore the i-f circuit must pass 189.2, 180.0, and 190.8 kc. The i-f channel frequency characteristics are shown in figure 7-13.

The effect of the Doppler shift, if the attacking ship is approaching the target, is to raise the frequency received. The intermediate frequency, however, is lowered by this Doppler because the incoming frequency with its Doppler shift is subtracted from the oscillator frequency to give the intermediate frequency.

  Frequency characteristics of the r-f and i-f
amplifier combination.
Figure 7-14 -Frequency characteristics of the r-f and i-f amplifier combination.

Figure 7-13 shows that the zero doppler condition is displaced to a center frequency of 190.75 kc, thus allowing the shift caused by a closing range rate of 39 knots to be passed with only a slight loss, and a 19.5-knot doppler to pass at full gain. If the doppler is higher than 39 knots, part of the lower side band is lost, and at 58 knots the lower side band is lost completely, resulting in a 2.5-db loss in signal-to-noise ratio. In the opening doppler condition, the upper side band is lost at 19

Phase-shifter and buffer circuits.
Figure 7-15 -Phase-shifter and buffer circuits.


Conjugate detector and envelope filter.
Figure 7-16 -Conjugate detector and envelope filter.
knots. Even with the 2.5-db loss, the signals are usually strong enough to permit satisfactory operation at 58 knots closing and 19 knots opening. The over-all characteristics of both the r-f and i-f stages are shown in figure 7-14.


It has been explained that the 240-to-250 kc signal from the oscillator is supplied to the sum circuit through a buffer and to the diff circuit through a phase shifter, in order to obtain the proper zero or 180° phase relation between the sum and diff signals.

The phase difference between the two channels (shown in figure 7-15) is obtained by shifting the phase of the 240-250 kc supply to the diff channel with respect to the supply to the sum channel. In order to compensate for phase variations between the two channels inherent in manufacture, a phase-shift compensation has been incorporated in the phasing circuit. The phase-shift network

  has three fixed taps corresponding to 0°, 90°, and 180°. The 0° and 180° points are connected to opposite ends of a potentiometer and the 90° point is connected to the center of the potentiometer. The output of the circuit is taken from the moving arm of the potentiometer. As the phasing potentiometer is rotated through its range, the phase of the input to the diff modulator shifts smoothly and continuously from 0° to 180° with respect to the 240-250 kc signal from the oscillator with about a 3-db change in amplitude. The network in conjunction with the phase potentiometer, inserts an average phase shift of 90° with ±90° available for compensating purposes.

The 240-to-250 kc supply is connected to the sum modulator through a cathode-follower buffer stage with essentially no gain or loss and zero phase shift. The purpose of the stage is to prevent crosstalk between the strong signal in the sum channel and the weak signal in the diff channel for the on-target condition.


Derivation of DDI voltage from sum and diff.
Figure 7-17 -Derivation of DDI voltage from sum and diff.


The primary purpose of the conjugate detector is to combine the sum and diff voltages so that an up voltage produces an up deflection on the cathode-ray tube, a down voltage produces a down deflection, and an on-target signal produces no deflection at all or equal deflections in both directions. Figure 7-16 illustrates the schematic circuit of the conjugate detector and envelope filter.

The circuit is arranged so that the (S+D) (sum +diff) voltage is peak-rectified in the upper section of V714 and the (S-D) voltage is peak-rectified in the lower section. The terminology "S+D" and "S-D" is arbitrary, and is used to indicate that the sum and diff signals add in one circuit and subtract in the other.

The following equations apply to an assumed transducer:

Esum-=NE cos θ/2

Ediff-=NE cos θ/2

The practical transducer used with this system differs somewhat from the theoretical transducer assumed in the sine and cosine relation. However, the performance is developed for the case assumed and then is compared with the practical case. The sine and cosine relation is shown graphically in figure 7-17, A, in which the peak amplitudes of the sum (S) and diff (D) curves are assumed to be equal, the equality being established by the RCG circuit, which will be discussed later. Also plotted is -D, which is used in determining (S-D).

Figure 7-17, B, shows a plot of (S+D) and (S-D) as the electrical phase between up and down is varied. This plot can be thought of as either (1) the peak a-c voltage applied to the rectifier (figure 7-16) without regard to phase, or (2) the d-c voltage after rectification. The -(S-D) curve is also shown, as it is used in determining the output of the conjugate detector. The (S+D) and (S-D) curves indicate what is actually measured with respect to ground on the leads marked "sum+diff" and "sum-diff" in figure 7-16.

However, the difference between these two voltages is used to deflect the spot on the cathode-ray tube. In order to produce the currents necessary for deflection, the voltage output of the

  conjugate detector is supplied to the vertical deflection coils of the magnetically deflected cathode-ray tube through vertical deflection amplifiers. If an on-target echo is received, the (S+D) and (S-D) voltages are equal, and two pulses of equal amplitude are sent to the deflection coils. The windings of the deflection coils are wound in such a manner that these currents oppose each other and the net flux produced is zero, so that the spot is not deflected. If an up signal is received the (S+D) voltage is always greater than the (S-D) voltage and the currents establish a field that deflects the spot in a manner that indicates to the operator the deviation from true depression angle. For a down signal the converse applies.

A plot of this function, as the up-down phase shift is varied, is shown in figure 7-17, C. because d-c potentials are used, a plus sign indicates a positive potential for an up deflection on the cathode-ray tube, and a minus sign indicates a negative potential for a down deflection on the cathode-ray tube. This plot is known as the DDI window, because it shows the window width through which the system is sensitive.

Zero potential is obtained for zero phase shift, which corresponds to an on-target signal. Zeros are obtained also for +180° and -180°, which are not on target but are approximately 4 mechanical degrees off-target. For the practical transducer, however, the probability of ambiguity is very small, as the amplitude of the echo at 4° or more off-target is approximately 15 or 20 db below the amplitude for on-target. It can be seen that the peak amplitude of the BDI deflection occurs at ±2 mechanical degrees.

For simplicity, the sum and diff voltages in the receivers, figure 7-17, A, are shown as being equal. In actual practice, however, the sum voltage is made approximately twice the diff voltage in order to improve performance. This relation is accomplished by constructing the output coupling transformer of the sum channel to have a voltage gain of 6 db higher than that of the diff channel. The curves for these operating conditions are shown in figure 7-17, D, E, and F. From a comparison of figure 7-17, C, and figure 7-17, F, the effects of increasing the sum voltage can be seen. The peak amplitude shown in figure 7-17, F, is at approximately 127 electrical degrees, rather than at 90°.


Effects of the threshold circuit on No Signal, Weak Signal and Strong Signal.
Figure 7-18 -Effects of the threshold circuit.

Simplified schematic of the RCG circuit.
Figure 7-19 -Simplified schematic of the RCG circuit.
The preceding discussion was on the basis of an ideal transducer. A practical transducer has a somewhat different performance. This performance is shown in figure 7-17, G, H, and I, plotted in mechanical rather than electrical degrees, The effects of the threshold-out and threshold-in curves will be explained later. Beyond ±5 mechanical degrees a reversal is obtained so that up indicates down, and vice versa. However, the sensitivity of the transducer beyond 5° from the axis, is so low that unless the echo-ranging conditions are exceptionally good the signals are lost in the ambient noise and reverberation.


In figure 7-16 a potentiometer is shown connected between the -105 volt supply and ground, with the moving contact supplying this variable voltage to the plates of the conjugate detector. This potentiometer is called the threshold control,

  and its purpose is to produce an apparently higher signal-to-noise ratio in the DDI. It accomplishes this function by placing a bias on the plates of the conjugate detector, variable in magnitude from 0 to approximately -105 volts. The control is preset at the factory to provide a bias of -85 volts. This bias is of such a value that most of the noise is not rectified, but the signal plus noise rides above the threshold voltage. The variable supply to the diode is bypassed to ground by a capacitor, to provide a low a-c impedance to ground, regardless of the setting of the potentiometer.

Previously, when the conjugate detector was discussed, the effect of noise was not considered. Figure 7-18 shows the (S+D) and (S-D) voltages, (S+D)-(S-D) , and the presentation on the cathode-ray tube. Figure 7-18, A, shows this combination for no signal present.

If no signal is present the noise and reverberation, for a particular instant of time during the receiving interval, are constant and substantially


Gain plot.
Figure 7-20. -Gain plot.

independent of the tilt of the transducer. This condition is shown on the (S+D) and (S-D) row of figure 7-18. The setting of the threshold potentiometer is at the proper level to allow the noise and reverberation to just break through the threshold voltage. Inasmuch as an echo is not being considered-but noise or reverberation, which has a random phase relationship among its many components-the (S+D)-(S-D) plot also has a random nature. Because of the action of the threshold bins, the chart shows only the

  combination of those parts of the (S+D) and (S-D) voltages which are greater than the threshold voltage.

For a weak signal from a target that is exactly on the axis of the transducer, it is possible that the resultant of (S+D) and -(S-D) might not exceed the threshold voltage, with the result that no indication will appear on the cathode-ray tube. This condition is not so serious as might be imagined, however, because most targets are not exactly at 0°, and if the indication is slightly off-target, the signal overrides the threshold voltage.

In figure 7-17, F, and I, it can be seen that the DDI voltage plot is altered when the threshold voltage is applied to the circuit. In the DDI voltage for the practical transducer (figure 7-17, I), the effect is to move the point of maximum voltage from about 2 degrees off-target to 1.3 degrees. It should be noted also in the threshold-out curves that there are secondary patterns existing beyond ± 5°. If the noise and reverberation conditions are good, these secondary patterns may give deflections in the wrong direction-that is, if an echo is 6° below the axis of the transducer, the cathode-ray tube indicates up rather than down. If the threshold control is all in, then from figure 7-17, H, it is apparent that the secondary pattern is not utilized.


Following the conjugate detector is the envelope filter network (figure 7-16). The purpose of this

Audio circuit schematic.
Figure 7-21. -Audio circuit.

Typical audio characteristics.
Figure 7-22 -Typical audio characteristics.

network is to pass the desired frequencies and to attenuate the undesired noise frequencies. In this case, the desired frequencies should be spoken of in terms of pulse length rather than frequency. The length of the pulse appearing at the output of the detector is determined by the length of the transmitter pulse, which, in this system, is between 5 and 50 milliseconds. If the proper R and C values are selected for the filter, pulses of shorter than 5 milliseconds length are attenuated. The 800-cps modulation of the output wave, which is necessary for audio reception, is not used for the cathode-ray presentation. In fact, the envelope filter has sufficient loss at 800 cycles per second, so that this frequency appears only as a fringe on top of the pulse-if it is not lost in the ambient noise.


The gains of both channels of the receiver unit are controlled by the RCG circuit, which operates

  from the peak amplitudes of the reverberations. For an echo-ranging receiver the maximum usable gain is desired at all times so that the weak signals-which are lost if the gain is too small-are amplified as much as possible for use in the loudspeaker or in the cathode-ray tube. However, if the gain is too great the ambient noise or reverberation overloads the receiver at some stage so that signals, normally stronger than the reverberation, also overload and are not recognized. By means of the RCG circuit, the receiver gain is adjusted automatically so that the reverberation is kept at substantially a constant level at the plate circuit of V705 and at approximately 18 db below overload.

The RCG action is similar to the AVC action except that the RCG action is not reversible-that is, the circuit is arranged so that the gain can increase from a very low initial condition during the receiving interval but cannot decrease because the low-gain condition is re-established during each transmitting interval.

Figure 7-19 shows a simplified schematic diagram of the RCG circuit. During the transmitting interval a potential of about -10 volts is applied to the grids of the control tubes of the i-f amplifiers. This potential reduces the transmission through the system by more than 120 db.

Figure 7-20 shows how the receiver unit varies in sensitivity because of the time variation of reverberation. Inasmuch as the RCG circuit always produces a constant output of reverberation, figure 7-20 also shows how the amplitude of reverberation varies with time. This plot can be shown only in a general manner, because the amplitude of reverberation is extremely variable.

The actual operation of the circuit is much the same as that in the dual-channel receiver described at the beginning of this chapter. One difference is that in the sum and diff receiver, the reverberation level in the diff channel is approximately 5 db less than that in the sum channel. Because the gains of the two channels must be equal or integrally related at all times, it is necessary that two separate RCG circuits be employed. The only circuit difference in the two channels is the positive bias supplied to the RCG detector. The sum channel detector is biased to approximately +15 volts while the diff channel detector is biased to approximately +4 volts. This results in a greater


RCG voltage supplied to the sum channel and preserves the relative amplitudes of the signals as they pass through the sum and diff channels.

This circuit also incorporates a TVG switch. Its effect when in the on position is to delay recovery of the receiver to about 440 yards. normally with the switch off the gain is restored to within 3 db of maximum in 100 yards.


The audio detector receives its excitation from the sum input to the conjugate detector. The schematic of the audio circuit is shown in figure 7-21. The detector operates in a conventional manner, and the output from it consists of the 800-cps modulation applied to the transmitter and the

  noise voltages received by the transducer. The filter at the output of the detector is sharply tuned to 800 cycles per second, giving considerable attenuation to the noise frequencies. Figure 7-22 shows a graph of the typical audio characteristics of the receiver.

A 1-stage audio amplifier of conventional design is used to amplify the signal to the necessary power level of about 2 watts.

For test purposes, provisions are made to deliver the output of the audio amplifier to a 10-ohm resistor instead of to the loudspeaker. A headset jack also is provided with a resistor in series with it to drop the power being supplied to the headset to a reasonable value.


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